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A Circularly Polarized 60 GHz Microstrip Antenna - …

A Circularly Polarized 60 GHzMicrostrip AntennaHere is a mm-wave Antenna suitable for WLAN systemsBy V. A. Volkov, M. D. Parnes Ascor, andV. D. Korolkov and R. G. ShifmanResonanceThis article presents thedesign of a Microstrip radia-tor-based Circularly polarizedantenna for the operation range of59 to 61 GHz. Characteristics of theantenna are appended. This anten-na, owing to its inherent simplicity,cheapness and small size, can beused in transceivers for 60 GHzwireless local and high-capacity in-office wireless local networks haverecently become in large addition, since the centimeterwaves range has been congested,international recommendationsappeared to bring some of the mil-limeter wavelength subranges tocommercial employment. Figure 1 shows the 60 GHz wireless local net-work that is intended for indoor use, allowingdata transfer at the rate of 10 Mbit/s.

A Circularly Polarized 60 GHz Microstrip Antenna Here is a mm-wave antenna suitable for WLAN systems By V. A. Volkov, M. D. Parnes Ascor, and V. D. Korolkov and R. G. Shifman

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Transcription of A Circularly Polarized 60 GHz Microstrip Antenna - …

1 A Circularly Polarized 60 GHzMicrostrip AntennaHere is a mm-wave Antenna suitable for WLAN systemsBy V. A. Volkov, M. D. Parnes Ascor, andV. D. Korolkov and R. G. ShifmanResonanceThis article presents thedesign of a Microstrip radia-tor-based Circularly polarizedantenna for the operation range of59 to 61 GHz. Characteristics of theantenna are appended. This anten-na, owing to its inherent simplicity,cheapness and small size, can beused in transceivers for 60 GHzwireless local and high-capacity in-office wireless local networks haverecently become in large addition, since the centimeterwaves range has been congested,international recommendationsappeared to bring some of the mil-limeter wavelength subranges tocommercial employment. Figure 1 shows the 60 GHz wireless local net-work that is intended for indoor use, allowingdata transfer at the rate of 10 Mbit/s.

2 Twomicrostrip antennas (receiving and transmit-ting) are connected to a high frequency antennas are built from a number ofstandard units, at the user station and from asingle unit, in the operation sites radiate and receive Circularly polarizedwaves to minimize the influence of multiplereflections in the room. In the operation sitesdistributor, the Antenna beamwidth from 60 to80 degrees would suffice to cover a rather article discusses the design sequence of asingle-unit Circularly Polarized 60 GHzmicrostrip Antenna . A Circularly Polarized elec-tromagnetic wave can be obtained using twomutually perpendicular, equi-amplitude, linearlypolarized waves that are in phase quadraturerelative to each other. Then, if a Microstrip patch(MSP a square-shaped Microstrip radiator) isexcited on two sides by equi-amplitude oscilla-tions with the phase difference of 90 degreesbetween them, the total field radiated by allMSPs will be Circularly idea of circular polarization resultingfrom the excitation of two spatially orthogonalmodes in the Antenna is the basis for all rotarypolarization MSPs.

3 Various approaches canmake the idea a reality. Presented here is thesimplest technique: a quadrature hybrid is usedto divide the power into two equi-amplitudeparts in phase and designFigure 2 illustrates the Antenna design andmethod of mating with the UG-358/U standard34 APPLIEDMICROWAVE& WIRELESS Figure 1. A 60 GHz wireless local APPLIEDMICROWAVE& WIRELESS waveguide flange. A board (1) (h= mm in thick-ness), including both the radiator and the bridge,together with the mm-thick dielectric layer (2) andthe upper screen (3), form a balanced stripline with b= mm. The Antenna pedestal is designed as a specialconfiguration waveguide flange (4) with a mm deepslot to receive both the strip circuit and the waveguide-to-stripline adapter. A quarter-wavelength waveguideshort (5) provides the optimal connection between thebalanced stripline and the standard-size waveguide ( mm).

4 The strip board incorporates the Microstrip patch(radiator) and the quadrature hybrid. In this case, theradiator and the matching quarter-wave transformerare constructed on an unbalanced Microstrip line, whilethe hybrid, the pill and the waveguide-to-strip adapterare based on a balanced stripline. The Rogers sheet dielectric (R3003,h= mm, e= ) was selected as material for the Antenna . First,both the geometry and the frequency response of themicrostrip part of Antenna were defined. Then the MSPwith the quadrature hybrid and calculation of polariza-tion characteristics of the Antenna was 3 depicts the Antenna configuration and thesix-pole equivalent circuit where 1 and 1' are the inputterminals and 2 and 2' and 3 and 3' are outputs of high-resistance probes. The electric models do not take intoaccount physical dimensions of the probes consideringthem as lumped elements.

5 With the supposition of neg-ligible spacing between exciting points compared to thefree-space wavelength, it is obvious that complex ampli-tudes of the resulting spatially-orthogonal electric fieldcomponents Exand Eyvary in proportion with the trans-mission gains s12and s13, respectively. That is, Ex~ s12and Ey~ s13or Px~ |s12|2 and Py~|s13|2, where Pxand Pyare the averaged densities of the power fluxestransferred by the orthogonal components Exand Ey,respectively).Accordingly, estimation of MSP polarization proper-ties is reduced to calculation of the full multiport scat-tering matrix, using the microwave device analysis pro-gram similar to that described in [1]. The program per-mits calculation of the full multiport scattering matrixfrom known s-matrices of its components. Whereas theprogram leaves room only for editing to allow new ele-ment definition subprograms to be entered. We have cor-rected the available description of the element the openend of Microstrip line, MSP, according to the termi-nology adopted in the is known that MSP can be modeled by the equiva-lent circuit in the form of a transmission line sectionwith the length that differs from the actual length Lbythe value l, which is the MSP extension on both ends,due to the edge capacitances (such as with allowancemade for existence of the edge fields).

6 The transmissionline is loaded by the real resistance Rrthat is equivalentto the ohmic loss by energy dispersion influence on the quasi-static expan-sion lstvalue was studied in [2], where the authorshave shown that, for the millimeter range of wave-lengths, the dispersion expansion lfappears to be sub-stantially shorter than the lstvalue obtainable using aquasi-static shown in [3] and [4], the frequency depended waveimpedance Z0(f) and the effective dielectric constant ere(f) can be represented as follows: (1)(2)where(3)GZZ= +00505600 rerrrepfGff()= + 1Zf ZZZG ffttp000021()= + Figure 2. Configuration of the Antenna . Figure 3. Printed-circuit board screen with openingQuarter-wavelength waveguide shortDielectic layerPrinted-circuit board with radiatorSpecial-purpose flange38 APPLIEDMICROWAVE& WIRELESS(4)In these equations, fpis measured in GHz, his mea-sured in millimeters, and Z0and Z0tare measured inohms.

7 Z0andZ0tare the wave impedance of the striplineof width (w), height (hand 2h), respectively. eeris theeffective dielectric constant. BothZ0and eerare theknown quasi-static values [1].In the final analysis, the input reflectivity of the L+ l(f) long regular line, when loaded by the real resis-tanceRr, was computed from the known formula withregard to imaginary inhomogenities and to the effect ofdispersion. A novel expression has been obtained for Rrfrom numerous experimental observations:(5)where(6)(7)(8)Figure 4 gives estimated frequency dependencies of s11 2for the Microstrip patch, mm in size,with the matching quarter-wave transformer, eitherwith or without regard to dispersion are two typical design alternatives for thequadrature hybrid. One topology provides a matchedload available in the fourth isolated arm; the other hasno load in the fourth arm. Figures 5 and 6 present both the gain-frequency andthe phase-frequency characteristics of the equivalentsix-pole circuit transfer coefficients s12and s13for bothalternative quadrature hybrid designs.

8 Lhffhhfrere=+ ++ 0 412030 2580 26408.() .() .. RZfer= ++ < 16090103500228100(),. for = 20ref()sjtg Ljtg LRRffrr11111111= +[]+ +()[] () fZhp=0 39760. Figure 5. The transfer coefficient gain- frequency (a) andphase frequency (b) characteristics of the equivalent six-pole circuit incorporating a matched load quadraturehybrid. Figure 4. Theoretical return loss for the Microstrip patch, mm in size, with the matching quarter-wavetransformer. Figure 6. The transfer coefficient gain-frequency (a) andphase-frequency of the equivalent six-pole circuit incor-porating a quadrature hybrid with no load. !" #$ $$ $ $ $ # !" #$ $$ $ Frequency, GHzIgnoring dispersionIncluding dispersionFrequency, GHzFrequency, GHzFrequency, GHzFrequency, GHz40 APPLIEDMICROWAVE& WIRELESSThe vital distinction between the show loaded circuitfrequency response and that of the unloaded circuit pro-vide clear evidence in support of the rather apparent butsometimes forgotten statement that any unreasonableidealization of the device to be developed is inadmissibleat the simulation resultsThe topology version presented in Figure 3 wasaccepted for implementation as the best suitable to fitrequirements of circular polarization.

9 It was supposed touse a dielectric wedge made of 16 m thick tantalum filmas a matched load in the bridge in Figure 7 is the MSP frequency characteris-tic that was measured with the following waveguide-to-strip adapter parameters: depth of the strip immersionin the waveguide the short circuit plane is tobe offset by the strip, where bis the widthof the narrow waveguide wall and l0is the waveguidewavelength that corresponds to the central frequency ofthe given range (60 GHz).Polarization characteristics were measured using ahorn rotating around the longitudinal axis. The measur-ing error due to the cross-polarization loss was about 20dB. The peak power response W was observed undersuch conditions when the horn plane Ewas parallel tothe major axis of the polarization ellipse and the mini-mum response W , when Ewas perpendicular to thataxis. Just as the amplitude ratio = E0x/E0y, so the phasedifference dbetween Exand Eywere determined fromthe experimentally found values of W , W and t(thepolarization inclination) using the relationships [5] thatcorrelate two forms of the polarization ellipse descrip-tion (canonic and parametric).

10 It follows then that apolarization ellipse can be represented parametricallyusing the field components and basing on the measuredvalues, as in the calculation: (9)(10)where E0xand E0yare amplitudes of orthogonal electricfield 8 shows both theoretical and experimentalpolarization ellipses for MSP radiated waves such as 59,60 and 61 GHz. Table 1 gives polarization ellipse para-meters, such as the axial ratio q, the inclination tandthe circular polarization efficiencypc, that were mea-sured with the frequency of 59, 60 or 61 GHz. The cir-cular polarization efficiency was determined from theexpression:EEtyy=+()08sin EEtxx=()0sin Figure 8. Polarization ellipses for MSP waves radiatedwith frequencies (a) GHz, (b) GHz and (c) $ $ # Figure 7. Antenna s return loss vs. ofFrequency, GHzPolarization596061q2, , degreed1515 Table 1. Polarization ellipse parameters measured at thefrequencies of 59, 60 or 61 of radius r r= Pmax= WmaxTheoryExperimentCircumference of radius r r= Pmax= WmaxTheoryExperimentTheoryExperimentFreq uency, GHz(a)(b)(c)Circumference of radius r r= Pmax= WmaxTheoryExperimentNOVEMBER1999 41(11) The Microstrip Antenna pattern isshown in Figure proposed Microstrip antennaprovides the required beam widthand circular polarization parametersat the frequency 60 GHz.


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